Quadrature networks are indispensable components used in virtually all kinds of radio frequency circuits. An ideal quadrature network is a three port device that produces from a single input two equal amplitude signals separated in phase by exactly 90 degrees. An ideal quadrature network 140 is illustrated in FIG. 1 where a sinusoidal input signal 100 is processed to produce an in-phase output 110 with a normalized amplitude that is 1/.sqroot.2 of the original signal amplitude and a quadrature-phase (hereafter "quadrature") signal 120 also 1/.sqroot.2 of the original signal amplitude which is shifter .pi./2 radians from the in-phase signal 110. The two output signals should have a relative phase shift of .pi./2 radians (90.degree.). Typically, quadrature networks produce an in-phase signal 110 that leads the input signal by .pi./4 and a quadrature signal 120 that lags the input signal 100 by .pi./4, i.e. for a total .pi./2 phase shift.
At microwave frequencies, quadrature networks may be realized by an arrangement of coupled transmission lines. Two examples 150, 160 of coupled transmission line quadrature networks are shown in FIG. 2. The size of these quadrature networks generally approaches that of a quarter wavelength .lambda./4 of the operating frequency and results in compact designs at microwave frequencies. Unfortunately, at UHF (e.g. 800 MHz), the large size of these coupled transmission line quadrature networks makes them extremely inconvenient for incorporation in relatively small radio units, e.g. hand held portable or mobile radiotelephones. Lacking the necessary ability to further miniaturize, transmission line quadrature networks are also not suitable for incorporation into Application Specific Intergrated Circuit (ASIC) devices.
One example of a compact form of quadrature network that may be used as part of an ASIC device is shown in FIG. 3. A so-called reactive quadrature network 300 is constructed from resistors and capacitors (as configured in the drawing) to realize a high-pass filter 334 and low-pass filter 332 combination. The selection of the 75 ohm resistors 325 and the 18 pF capacitors 320 produces a crossover frequency Fc (in this example) of approximately 116 MHz. The 50 ohm input and output resistors 330 provide impedance matching. Obviously, other resistor and capacitor values would be used to provide operation at different frequencies.
The function of quadrature network 300 is illustrated by output signal phase 340 and output signal amplitude 350 graphs shown in FIGS. 4(a) and 4(b), respectively. As indicated by the graph of output phase 340, the in-phase output 310 leads input signal 305 by 45.degree., and the quadrature signal 325 lags the input signal by an equal amount. This results in a 90.degree., or quadrature, relationship between output signals that is essentially independent of frequency. The high-pass/low-pass nature of the quadrature network 300 is illustrated by the output signal amplitude graph 350. The in-phase output 310 is connected to the high-pass filter 334, and the in-phase output 310 increases in signal amplitude with increasing frequency. Conversely, the quadrature output 315 is connected to the low-pass filter 332 and increases in signal amplitude with decreasing frequency. At the crossover frequency, Fc, the signal amplitudes of both the in-phase and quadrature signals are equivalent, and the quadrature network acts as a power splitter with approximately 5 dB of resistive loss in addition to the 3 dB power split. In order words, each output signal contains exactly half of the total output power.
Although the relative quadrature phase difference (i.e. 90.degree.) is essentially frequency independent as can be seen in FIG. 4(a), the power split is strongly a function of frequency. For example, with increased frequency beyond F.sub.c, the in-phase output is at a higher power than the quadrature output. It is difficult in practice to achieve both optimal quadrature performance and an equal power split. In addition to these frequency dependent effects, inequalities due to process spread, temperature, etc. in the resistors 325 and capacitors 320 also result in imbalances to the quadrature phase difference and power split.
Another type of compact quadrature network is a doubler-divider circuit where a mixer or other non-linear device is used to double the frequency of the input signal. Two frequency dividers then reproduce the original frequency using the positive and negative slopes of the doubled signal to generate the 90 degree offset. This technique is somewhat limited depending on frequency range because of the requirement that the RF signal be doubled. For example, at 800 MHz the double-divider circuit must operate at 1.6 GHz which exceeds the operating speeds of most silicon ASICs.
Quadrature networks are often used in the modulation and demodulation of voice communication signals. Modulation is the process by which the information content of an audio signal is transferred to an RF carrier for transmission. The inverse process (i.e., recovering the audio information from an RF signal) is called demodulation. The modulation process causes some property, such as the amplitude, frequency, or phase, of a high-frequency carrier to be deviated from its unmodulated value by an amount equal to the instantaneous value of the modulation (e.g. voice) signal. There are many forms of modulation/demodulation of which amplitude modulation, (AM), and frequency modulation, (FM), are perhaps the most well know. More complex modulation techniques which are often used with various digital signaling schemes require the use of a quadrature modulator. Quadrature modulation is advantageous (relative to purely AM and FM) in reducing spectral occupancy, obtaining voice privacy, reducing envelope fluctuation, and reducing transmit power demands.
In digital communications, all quadrature modulation techniques begin with an analog voice signal being sampled and converted into a digital data stream. In the simplest form of quadrature modulation, quadrature phase shift keying (QPSK), the audio information is separated into two individual digital data streams: the I, for in-phase digital data stream 425, and Q, for quadrature digital data stream 430, as shown in FIG. 5(a). Each of these data streams is individually mixed in respective mixers 405 and 410 with a local oscillator signal 435 that has been passed through a quadrature local oscillator network 420. The "ideal" quadrature network 420 splits the signal from local oscillator 435 into two equal amplitude signals at the same carrier frequency but separated in phase by exactly 90.degree.. The in-phase signal I 425 is mixed with the in-phase cosine wave from network 42 and the quadrature signal Q 430 is mixed with the quadrature sine wave from network 420. The two modulated signals (i.e. the I and Q information two modulated signals (i.e. the I and Q information modulate the local oscillator carriers) output from mixers 405 and 410 are combined in summer 415 to produce a single QPSK modulated output signal.
On the receiving end, an ideal QPSK demodulator 450 shown in FIG. 5(b) recovers the original data streams. The QPSK demodulator 450 first splits the received input signal into two equal parts which are separately fed into the in-phase mixer 460 and the quadrature mixer 465. The mixers are fed with quadrature signals provided by the "ideal" quadrature network 470. After some post-mixing filtering (not shown), the individual I and Q digital data streams 475 and 480 are recovered.
QPSK modulation produces a four point constellation 485 in the complex power domain (sometimes referred to as a phase diagram) shown in FIG. 5(c). QPSK modulation causes the QPSK modulator 400 output signal, which shifts between the values (0,0), (0,1), (1,0), and (1,1) depending on the I and Q channel values, to occasionally swing through zero output power. The resulting envelope fluctuation between zero and some maximum output power requires the use of a low-efficiency linear power amplifier to avoid distortion and spectral spreading.
Envelope fluctuations and their deleterious effects can be reduced by using a more complicated form of quadrature modulation known as .pi./4-shifted quadrature phase shift keying or .pi./4-shifted APSK. FIG. 6(a) shows a .pi./4-shifted QPSK modulator that operates exactly as the QPSK modulator 400 described in conjunction with FIG. 5(a) except that the output signal is alternately shifted by .pi./4 radians. This shift is not shown in the figure because it is preferably accomplished computationally by adding a .pi./4 shift to the I and Q bit streams on alternate cycles using for example a digital signal processor (DSP). This .pi./4-shifted QPSK output produces the constellation 545 shown in FIG. 6(b) where it may be seen that zero output power crossings are avoided, thereby permitting the use of high efficiency nonlinear power amplifiers. To achieve this benefit, the IS-54 standard for North American digital cellular communications specifies that the modulation be based on the differential phase rather than the absolute phase of the received signal, defined above as .pi./4-shifted DQPSK.
Unfortunately, in practical quadrature networks (as opposed to the ideal quadrature networks described above), amplitude and phase imbalances in those networks distort the shape of the transmitted or received constellation. As the constellation of the modulated output is distorted, it becomes increasingly more difficult to correctly identify the actual phase of the received signal. Errors in phase identification result in increased bit errors in the I and Q data streams. As the number and frequency of these bit errors increase, the overall quality of the reconstructed voice signal is reduced. Since voice quality is a primary goal of any voice communication system, it is essential that phase and amplitude errors in the quadrature networks of these modulators be reduced.
Quadrature networks also find application in single side-band (SSB) upconverters. Upconversion is the process by which two sinusoidal signals are multiplied together (i.e. mixed), to produce a frequency which is the sum (upper sideband) or difference (lower sideband) of the constituent frequencies. Referring to the SSB upconverter 600 shown in FIG. 7, an intermediate frequency (IF) oscillator signal 610 (the signal to be "upconverted") is passed through a quadrature network 620 whose outputs are fed into two separate mixers 660, 650. An output signal from RF oscillator 630 is passed through a quadrature network 640 whose outputs are also fed into the mixers 660, 650 as shown. The difference frequencies (i.e. the lower sidebands) generated by each of these mixers are fed into the summing junction 670 (ideally) 180 degrees out of phase with one another. Accordingly, the lower sidebands cancel each other and are thereby eliminated. An advantage of SSB upconverters is that they do not require the use of a sharp cutoff filter to remove the difference frequency. A disadvantage of SSB is that any phase or amplitude imbalance in either of the quadrature networks 620, 640 results in incomplete cancellation of the difference frequency thereby producing unwanted spurious outputs.
The present invention provides an improved quadrature network that substantially reduces the sensitivity to amplitude and phase errors in generating quadrature LO signals. First and second single side band (SSB) networks receive first and second input signals. The first SSB network mixes in-phase components of the first and second input signals and quadrature components of the first and second input signals and combines these first mixed signals to generate an in-phase signal. The second SSB network selectively mixes in-phase and quadrature components of the input signals and combines these selectively mixed signals to generate a quadrature signal. The thus generated in-phase and quadrature signals are equal in amplitude and out of phase by 90.degree.. Any amplitude or phase imbalance between any of the in-phase and quadrature components is substantially eliminated in the quadrature network. As a result, the quadrature local oscillator network according to the present invention is insensitive to component manufacturing tolerances and temperature effects, and is particularly useful in quadrature modulators and demodulators used in mobile/portable voice communication systems both in radiotelephones and in base stations.